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Frontiers in Electronics ·

Volume 7 - 2026 | https://doi.org/10.3389/felec.2026.1939862

Research on capacitive radio frequency non-contact communication devices under magnetic resonant power transfer

  • Shuyao Tian1
  • Chong Zhang1
  • Xiaoqiang Wu1
  • Pengyan Yao1
  • Yangyang Xiang1
  • Hanzhi Zhao1*corresponding author
  1. School of Electronics and Control Engineering, North China Institute of Aerospace Engineering, Langfang, Hebei Province, China
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Abstract

Common industrial rotating electrical connection components often utilize contact-based connectors such as slip rings. However, during prolonged operation, issues like poor contact, increased friction, or even fires may arise. Non - contact power and data transmission technologies offer a promising approach to mitigate these problems, although their long-term dynamic performance requires further evaluation. Nevertheless, traditional wireless communication technologies exhibit instability and insufficient interference resistance in high-voltage enclosed environments. Addressing this, the paper systematically analyzes the characteristics of capacitive radio frequency (RF) transmission in a setup based on parallel-parallel magnetic coupling resonance and capacitive RF technology. Through simulation studies, the impact of the installation position of the wireless power transmission module and the isolation material on the performance of capacitive RF data transmission is investigated. A non-contact capacitive RF data channel with magnetic coupling resonance magnetic field isolation capability is designed. Based on this design, a non-contact magnetic coupling power and capacitive RF data co-transmission experimental platform with a rated capacity of 3 W is developed, verifying the feasibility and effectiveness of the proposed scheme. Simulation results indicate a peak baseline transfer efficiency of 84.53% under scaled-down conditions (492.2 mW), while experimental platform testing at the 3 W rated power achieves a transmission efficiency of 81.2%. Moreover, using level buffer amplification, the experimental data transmission rate reaches 14.4 kB/s (net effective data rate, corresponding to an uncoded raw bit rate of 115.2 kbps) with a bit error rate below 10 −6 on a stationary test bench. Although dynamic downhole verification under high-speed rotation and vibration remains for future work, this hybrid topology demonstrates the baseline feasibility of simultaneous, isolated power and high-speed data transmission between embedded drilling modules.

Keywords

1 Introduction

In the field of petroleum drilling, various application scenarios such as near - bit geosteering (Zhdaneev et al., 2021), measurement while drilling (MWD) (Kück et al., 2021), and downhole fish imaging recognition frequently require short-distance data and power transmission. Traditional methods involve the use of slip rings or brushes at the electrical connections of rotating drilling rigs. During drilling operations, conditions typically include high temperatures and triaxial high-frequency strong vibrations. Due to the manufacturing processes and mechanical relative motion of slip rings and brushes, issues like arcing, burns, and localized heating are prone to occur during drilling rotations (Tomiyoshi et al., 2025; Zhang et al., 2024; Zhao et al., 2025), significantly reducing the reliability of electrical connections. Installing high-temperature-resistant batteries in embedded drilling systems to power downhole circuits can reduce frictional contact surfaces (Mansour et al., 2025; Ye and Parspour, 2024), but they cannot sustain long-term operation, thus incurring additional unnecessary maintenance costs. Moreover, in the event of high - temperature explosion or combustion failures, substantial and difficult-to-estimate damage can be caused to the embedded drilling circuits (Afonso and Liu, 2024; Djidjekh and Takacs, 2025; Tsai and Liu, 2025).

Consequently, wireless power transmission can effectively address power disconnection issues caused by installation errors and mechanical vibrations. Although wireless communication technologies such as Long Range Wide Area Network (LoRa) (Shang et al., 2025; Soto-Vergel et al., 2023), Bluetooth (Hanifi et al., 2025; Z. Li et al., 2025; Top and Gökbulut, 2022), ZigBee (Yuan et al., 2021), mobile communications (Chen et al., 2025), RFID (Radio Frequency Identification) (Zhao et al., 2024), and acoustic communication (Stojanović et al., 2025) are capable of facilitating data transmission for embedded sensors, they are not suitable for use in the harsh downhole environment characterized by strong vibrations and high temperatures. Consequently, the development of an efficient and isolated transmission system for both power and data holds significant practical value. The current predominant methods of wireless power transmission (WPT) include microwave radiation (Razek, 2021), electromagnetic induction, ultrasonic, and magnetic resonant coupling (Agunloye, 2023). Microwave radiation offers the advantages of long transmission distances and strong directivity, but it requires continuous alignment of the microwave direction. In contrast, magnetic resonant coupling utilizes the energy-storing properties of the non-radiative electromagnetic near-field to harvest the evanescent waves emitted by high-frequency power sources, enabling wireless power transmission. Moreover, compared to traditional loosely coupled systems, magnetic resonant coupling can achieve reliable power transmission over relatively greater distances.

In the coupled resonant energy and data modulation transmission technology approach (Xu Y et al., 2024) proposes a wireless power transmission system based on power carrier modulation to simultaneously acquire temperature sensor data from a motor rotor. Meanwhile (Debela and Singh, 2021; Xu S et al., 2024), have developed and constructed a prototype for wireless charging feedback control in electric vehicles. However, the prototype is limited by the operating temperature and size constraints, which make it challenging to meet the requirements of downhole drilling measurements.

For data isolation transmission, common techniques applied in planar circuits include optical coupling isolation (Feng et al., 2024), magnetic isolation (Laha et al., 2023), and capacitive isolation (Mustapha and Sanduleanu, 2023; Wang et al., 2023). However, these conventional digital isolation devices are inadequate for isolation transmission loops subject to mechanical movement between different planes. Similar to the principle of coupled WPT, which uses alternating electric fields as an energy carrier to achieve wireless power transmission through coupled plates, this method ensures stable energy transfer even in the presence of metal objects between the coupled plates. Additionally, because the electric field is primarily confined between the coupled plates, the electromagnetic interference to surrounding transmission links and components is relatively minimal (Lindeman et al., 2023). Reference (Xu et al., 2025) designed a sleeve-type non-contact electric field coupling power transmission framework for rotating machinery. However, this coupling structure is prone to isolation breakdown and short-circuit faults under downhole severe vibration. Recent advancements in simultaneous wireless power and data transfer (SWPDT) have predominantly focused on inductive-inductive or inductive-NFC couplings (Feng et al., 2024). However, cross-talk between high-power magnetic resonant fields and sensitive inductive communication channels often restricts SNR and throughput. While capacitive data links offer higher bandwidth, existing planar geometries are vulnerable to downhole misalignments and magnetic field escape.

To overcome this cross-coupling bottleneck, this paper investigates a hybrid high-speed transmission architecture combining parallel-parallel (PP) magnetic resonant power transfer with a capacitive RF differential data channel in a non-planar parallel configuration, as illustrated in Figure 1. Through nanocrystalline magnetic shielding and spatial non-coplanar layout, the magnetic-to-electric cross-interference is significantly attenuated. A stationary prototype is developed to validate the feasibility of this isolated dual-channel transmission while identifying the structural and dynamic constraints for subsequent downhole packaging.

Structure of the proposed capacitive-RF simultaneous power-and-data transmission system.
Figure 1. Structure of the proposed capacitive-RF simultaneous power-and-data transmission system.

2 System theoretical modelling and analysis

2.1 Isolated wireless energy transmission implementation

To enhance the overall transmission efficiency of the system, compensation networks are often introduced to offset reactive power, thereby rapidly establishing a resonant network. Based on the differences between the compensation structures at the WPT transmitter and receiver ends, four common compensation methods are series-series (SS), series-parallel (SP), parallel-series (PS), and parallel-parallel (PP) (Costanzo et al., 2023; Silge and Sattel, 2018). Series resonant topologies offer simple control and can output higher power with a lower amplitude AC voltage source. This paper focuses on the working principle of the parallel-parallel (PP) topology, which is illustrated in Figure 2.

Equivalent circuit of parallel-parallel (P-P) topology.
Figure 2. Equivalent circuit of parallel-parallel (P-P) topology.

In the equivalent circuit (Figure 2), U s represents the fundamental RMS AC voltage input to the resonant module from the external full-bridge inverter circuit, operating at angular frequency ω = 2 π f. Q *1* - Q *4* are high-frequency inverter-driven power IGBTs and D *1* - D *4* constitute the diode rectifier bridge. C s and C r denote the parallel compensation capacitors on the primary and secondary sides, respectively, with R C s and R C r being their internal parasitic equivalent series resistances (ESR). L 1 and L 2 are the self-inductances of the transmitting and receiving coils, respectively, where R 1 and R 2 represent the high-frequency AC parasitic resistances of the coils. M is the mutual inductance between L 1 and L 2, and R L e q is the equivalent DC-side resistive load. Here, I 1 and I 2 represent the loop currents flowing through the primary coil L 1 and the secondary coil L 2, respectively, while I s is the total primary input current and I L e q is the current delivered to the equivalent load R L e q.

Ignoring the loss of external connecting transmission lines and the built-in parasitic parameters of the resonance compensation capacitors, the parameters of the electrical power transmission module along the radial axis of the drilling shaft are highly symmetrical, thus simplifying the calculation parameters. As analyzed in Figure 2, the equivalent impedance of the secondary winding is expressed in Equation 1:

Z 2 r e q = R 2 + j ω r L 2 + Z C r R L Z C r + R L (1)

Based on the definition and inference of mutual inductance between coupled coils, the equivalent impedance of the secondary receiving coil on the drilling side, reflected and converted to the primary coil, can be determined as shown in Equation 2:

Z 2 r e q ′ = M L 2 2 R L − j ω r M 2 L 2 (2)

Under these conditions, the analysis process of the primary side circuit can be simplified. The expression for the total impedance on the primary side is:where

Z 1 i n = a 1 + j ω r a 2 − a 3 a 4 + a 5 (3)

a 1 = R 1 + M L 2 2 a 2 = 1 − ω r 2 L 1 − M 2 L 2 a 3 = C s R 1 + R L M L 2 2 2 a 4 = 1 − ω r 2 L 1 − M 2 L 2 C s 2 a 5 = ω r 2 C s 2 R 1 + R L M L 2 2 2

When resonance occurs, the imaginary part of the corresponding impedance becomes zero. Referring to (Equation 3), the equivalent impedance of the circuit topology is simplified. After substituting and simplifying the operations, the currents I s, I 1, I 2, I L, as shown in Figure 1, can be determined (Equation 4):where Z 1 ′ represents the impedance of the transmitting end winding, and Z 2 ′ denotes the equivalent impedance of the receiving end with the load. By further derivation using (Equation 4), the primary coil’s transmission power P s and the secondary coil’s output power P o can be obtained, as shown in Equations 5, 6:

I s = j ω r C s U s + I 1 I 1 = U s Z 2 ′ Z 1 ′ Z 2 ′ + ω r 2 M 2 I 2 = − j ω r U s M Z 1 ′ Z 2 ′ + ω r 2 M 2 I L = − j ω r U s M 1 + j ω r C r R L Z 1 ′ Z 2 ′ + ω r 2 M 2 (4)

P 1 = U s 2 Z 2 ′ Z 1 ′ Z 2 ′ + ω r 2 M 2 (5)

P o = ω r M U s 2 1 + ω r 2 C r 2 R L 2 R L Z 1 ′ Z 2 ′ + ω r 2 M 2 1 + j ω r C r R L 2 (6)

Analyzing Equations 5, 6, it is evident that under a selected resonant frequency, the main parameters affecting the efficiency of power transmission include the mutual inductance M, the coil inductance L, and the output load R L. The output efficiency is directly proportional to the mutual inductance.

In the literature (Li et al., 2023; Song et al., 2019), for two coils in a coaxial arrangement, the mutual inductance M can be determined using the Neumann formula:where l 1 and l 2 represent the circumferences of a single turn in the primary and secondary coils, respectively. When the coil shape is circular, let N 1 and N 2 be the number of turns on each side. The vectors d l 1 and d l 2 are arbitrary position vectors on the loops l 1 and l 2, respectively, and θ is the angle between the two line elements in the polar coordinate system at their positions. A schematic of the coil positions is shown in Figure 3 (Shah et al., 2024), where r 1 and r 2 are the radii of the two coaxial coils, r is the geometric distance between the two vector elements, and h is the parallel spacing between the two coils.

M = μ 0 N 1 N 2 4 π ∮ l 1 ∮ l 2 d l 2 → · d l 1 → r (7)

Mutual inductance model of concentric parallel planar coils with axial offset.
Figure 3. Mutual inductance model of concentric parallel planar coils with axial offset.

By deriving from the mathematical relationship of the lengths of the sides of a triangle, the algebraic expressions for the product of the vectors d l 1 and d l 2 and their geometric distance r, as shown in Equations 8–10, can be obtained:

d l 2 → · d l 1 → = r 1 r 2 ⁡ cos ⁡ θ d θ d φ (8)

r = r 1 + r 2 2 + h 2 1 − 4 r 1 r 2 r 1 + r 2 2 + h 2 cos 2 ⁡ α (9)

t = 4 r 1 r 2 r 1 + r 2 2 + h 2 (10)

Simultaneously substituting into (Equation 7) and simplifying after variable replacement, a deduction from the Neumann formula involving complete elliptic integrals of the first and second kinds can be obtained. This deduction indicates that the main factors influencing mutual inductance are the radius of the primary coil r 1, the radius of the receiving coil r 2, and the spacing between the two coils h. By letting ϕ = r 1 / r 2 and ϕ 1 = h / r 2, the relationship between mutual inductance and the two ratios can be simplified and discussed separately. When the ratio of coil spacing to the radius of the receiving winding is fixed, calculations reveal a curve depicting how mutual inductance varies with the winding radius ratio, as shown in Figure 4.

**(a)** Trend of mutual inductance and pitch ratio under different coil spacing **(b)** Trend of mutual inductance caused by the ratio of coil spacing radius.
Figure 4. (a) Trend of mutual inductance and pitch ratio under different coil spacing (b) Trend of mutual inductance caused by the ratio of coil spacing radius.

Once the coil spacing h is fixed, there exists a specific ratio of the coil radii, ϕ = ϕ m, at which the mutual inductance M attains a maximum value, M max. As illustrated in Figure 4, the maximum effect is particularly pronounced when the ratio of the spacing h to the radius of the secondary coil, ϕ, is less than 0.8. Consequently, when determining the coil placement spacing, selecting a secondary coil radius r 2 that is greater than or equal to the coil spacing h will significantly enhance the mutual inductance M. When the radii of the transmitting and receiving coils are determined, altering the spacing h between the mutual inductance coils, as analyzed in Figure 3b, shows that regardless of the difference in radii between the coils, the mutual inductance monotonically decreases as the spacing between the coils increases. However, the rate of decay varies for different radius differences.

2.2 Isolated capacitive RF data transmission implementation

Data transmission is facilitated by two pairs of opposing planar metal capacitor plates, including a transmitting end and a signal receiving end, forming a coupled capacitor. This arrangement constitutes a signal transmission loop. Figure 5 illustrates the equivalent plate coupling topology applicable to coplanar annular planar capacitors.

Equivalent coupling topology between transmissive plates.
Figure 5. Equivalent coupling topology between transmissive plates.

Considering the coupling mechanism as a two-port network, where the transmitting and receiving plates can be regarded as the input and output of the network respectively, an intrinsic relationship can be derived to form a lumped parameter model. When an equivalent output load is absent, the port impedance can be determined using (Equation 11).

Z R F e q = 1 2 R P + Z x + 1 Z C P A B + Z y Z C P C D Z y + Z C P C D − 1 (11)

In this equation, Z x and Z y represent the equivalent impedances between the two opposing capacitor plates. Z C P A B, Z C P C D denote the internal resistances of the terminal capacitors, while R P is the internal resistance between the input and output lines. When using planar plates to establish an isolation capacitor, its capacitance value can be easily determined from the definition of planar plate capacitance. For different voltage levels of isolation standards, the selected plate spacing has a negative correlation with the capacitance value of the isolation capacitor. To improve the signal-to-noise ratio, a differential amplifier circuit is employed at the signal receiving end. The digital level output from the preceding stage system is transmitted to the receiving end in RF form via a power amplifier circuit. It passes through a pair of differential capacitor plates and the air gap between the plates, and after filtering, it finally enters the differential amplifier circuit’s non-inverting and inverting terminals. Additionally, it provides a return path for the input bias current of the operational amplifier. The output structure is depicted in Figure 6a.

Schematic of capacitive RF data transmission system. **(a)** Block diagram of capacitive data transmission principle; **(b)** equivalent noise model induced by component tolerance errors.
Figure 6. Schematic of capacitive RF data transmission system. (a) Block diagram of capacitive data transmission principle; (b) equivalent noise model induced by component tolerance errors.

The gain of the capacitive radiofrequency plate section for the transmitted signal is A C a p. According to the system’s equivalent model, the relationship between the signal gain and the input impedance at the receiving end is expressed in (Equation 12).where Z i n _ d i f represents the equivalent differential input impedance at the operational amplifier’s input terminal. Analysis reveals that the signal amplitude at the data receiving end is positively correlated with the gain. The signal gain ratio is influenced by the signal’s angular frequency, the equivalent capacitance of the RF capacitor C R F and the differential input impedance Z i n _ d i f.

A C a p = Z i n _ d i f Z R F e q + Z i n _ d i f = 2 R 1 Z R F e q + 2 R 1 (12)

To enhance the signal - to - noise ratio of the demodulation circuit, two filtering strategies are employed: pre - input signal filtering and post-differential to single - ended output filtering. Focusing on the former, a high - pass filter circuit is added before the input stage of the differential amplifier circuit, capitalizing on the frequency discrepancy between the operating frequency of the radiofrequency energy transmission module and the data transmission frequency. Consequently, the signal between the two capacitor plates can be considered approximately as a square wave signal source with resonant frequency interference. As depicted in Figure 6b, V c o e f denotes the interference source introduced by the resonant operating frequency of the radiofrequency energy transmission module, V s i g n a l represents the actual capacitive radiofrequency transmission voltage signal, C C M is the matching capacitor of the two branch filters, and Δ C is the capacitor introduction error. In Figure 6b, e n represents the common-mode noise at the input stage of the filter, while e t is the additional thermal noise interference introduced by different resistor values. Equations 13, 14 express the gain error voltage after the common - mode noise is differentially amplified in the circuit.

V i n P = V C M + V s i g n a l / 2 + e n j ω t R H f 1 + e t C C M + Δ C 1 + j ω t R H f 1 + e t C C M + Δ C (13)

V i n N = V C M − V s i g n a l / 2 + e n j ω t R H f 1 + e t C C M − Δ C 1 + j ω t R H f 1 + e t C C M − Δ C (14)

In conclusion, the introduction of a filter circuit before the differential amplifier stage can still result in additional common-mode interference due to differences in branch cutoff frequencies and resistance, even under ideal filter transition band conditions. To minimize the distortion of the output signal, it is necessary to calculate the amplitude of the received signal (with interference) and the common-mode input range of the selected operational amplifier, while applying isolation shielding to the radio energy transmission coil. It is preferable to choose an operational amplifier with a smaller input bias current to reduce the impact of increased input matching resistance on the output bias. The input stage signal should be amplified with a lower gain and then subjected to high-pass filtering to minimize the impact of radio energy transmission harmonics on the transmission signal as much as possible.

3 Simulation analysis

3.1 Output characteristics of wireless energy transmission modules

Considering the working and installation environment of the embedded power data transmission system during drilling, a symmetrical structure is chosen. Based on the conclusions mentioned earlier, the coil spacing is selected to be less than the radius of the secondary power receiving coil. Additionally, there exists a maximum mutual inductance coefficient radius ratio for a fixed spacing. Finite element simulations are conducted to validate and compare the mutual inductance coefficients for different coil spacings using solid copper enameled wire, as shown in Figure 1. The parameters of the simulation test platform are provided in Tables 1,2 presents the self-inductance and mutual inductance in air without the addition of nanocrystalline magnetic barrier materials.

TABLE 1. Parameters of transmission coil simulation.

ParameterValue
Primary coil radius, r *1* (mm)14
Secondary coil radius, r *2* (mm)14
Primary turns, N115
Secondary turns, N215
Wire diameter (mm)0.5
Primary input voltage, U *s*(V)15
Target resonant frequency, f (kHz)600
Equivalent load resistance, R L (Ω)250

TABLE 2. Simulation results of different axial spacing.

Parameter3 mm5 mm9 mm
Primary self-inductance, L1 (μH)11.918911.918811.9187
Secondary self-inductance, L2 (μH)11.930911.935511.9411
Mutual inductance, M (μH)6.61675.39584.4582
Coupling coefficient, k0.55490.45230.3737

According to Equations 7–9 mentioned earlier, the output power of the secondary receiving coil is primarily determined by the mutual inductance between the coils and the equivalent load resistance on the load side. As indicated by (Equation 10), the transmission efficiency and power of the radiofrequency energy transmission module are sensitive to the selection of the resonant frequency. In practical applications, due to installation errors and variations in material and craftsmanship, there is a positive correlation between the module parameter errors and the axial coil spacing.

For non-contact power transmission modules utilizing a parallel-parallel topology, the resonant circuit composed of an inductance-capacitance loop typically exhibits a relatively narrow band due to its high-quality factor. Under varying load conditions, establishing a compensation relationship between the load current and the primary coil current using Equation 7 can be relatively complex.

In an ideal state, the rectification transformation circuit and the subsequent shaping filter circuit in a wireless power transmission system with parallel compensation capacitors can be approximately equivalent to a load resistance. When the rectifier bridge conducts, the load of the resonant circuit undergoes nonlinear changes. Because the rectified signal contains a significant number of harmonic components, the introduction of load fluctuations based on this can lead to a shift in the overall resonant frequency.

The simulation test parameters for the excitation source are referenced from Table 1, and the parameters for the resonant mutual inductance coil are taken from the 5 mm entry in Table 2. Applying the fundamental approximation simplification method, which disregards the impact of higher-order harmonics, the output voltage of the compensation network can be approximated as its fundamental component, with the fundamental voltage and current always in phase. Consequently, the rectifier filter network can be considered as an equivalent purely resistive load. With an output filter capacitor value of 6.8 uF selected and a DC-side equivalent load resistance set to 150 Ω, the simulation data for voltage and current at the load resistance terminal are presented in Figure 7.

Load resistor voltage and current waveform after adding rectifier bridge: **(a)** current waveform; **(b)** voltage waveform.
Figure 7. Load resistor voltage and current waveform after adding rectifier bridge: (a) current waveform; (b) voltage waveform.

The current in the secondary resonant capacitor is ahead of the voltage, and when the output of the radio energy transmission module is applied, the distortion occurs, and then the resonance deviation occurs.

According to the spacing of 5 mm in Table 2, the electric field circuit co-simulation is carried out, and the RMS value of applied power can be calculated to show that when the load is 150 Ω, the transmitting power is 492.2 mW, the load resistance power is 416.02 mW, and the transmission efficiency is 84.53%. The simulation waveforms verify that the power requirements of embedded devices can be met by the voltage regulator loop at the load end using the proposed topology. To account for component tolerances and measurement uncertainties, repeated simulation runs indicate an efficiency variance of approximately ± 1.5%, confirming the stability of the 84.53% nominal efficiency.

It should be noted that the initial 492.2 mW simulation serves as a low-excitation baseline to characterize the theoretical upper limit of the resonant network without thermal non-linearities (yielding 84.53% efficiency). In the practical experimental setup, the input voltage and component specifications are scaled up to support the 3 W rated power required by downhole sensors, yielding a measured transmission efficiency of 81.2%.

This 3.33% efficiency discrepancy is attributed to: (1) increased I 2 R ohmic conduction losses in the solid enameled copper windings under higher excitation currents; (2) non-linear forward voltage drops and switching losses within the FGH40N60SMD IGBTs and SS310 Schottky bridges; and (3) slight magnetic core permeability dispersion and leakage in the nanocrystalline shield under elevated high-frequency field intensities. Readers should distinguish the theoretical low-power simulation baseline from the practical 3 W experimental efficiency. To ensure reproducibility, the capacitive RF plates are constructed from copper with a diameter of 30 mm and a thickness of 1 mm, and the magnetic shielding utilizes a nanocrystalline ribbon with a relative permeability μ r ≈ 10 4 and a thickness of 0.1 mm.

3.2 Effect of resonant position relative to capacitive RF pole plate on data transmission

For planar plate capacitors using air as the dielectric medium, their capacitance is quite sensitive to changes in spacing due to limitations in the dielectric constant and relative installation area. To ensure the reliability of the capacitive radiofrequency signal transmission path, the plates are typically maintained at a small distance apart. In contrast, the magnetic coupling resonant radiofrequency energy transmission coils essentially function as inductors operating at a higher frequency.

When operating at the resonant frequency, they generate alternating high-frequency magnetic fields around them, which can be approximately equivalent to radiating electromagnetic waves into the surrounding space. This coupling with surrounding conductive devices results in induced currents when absorbed by metal conductors, thereby affecting the capacitive plates used for high-rate signal transmission on the exterior. Figure 8 illustrates the coupling magnetic field distribution between the non-contact radiofrequency energy transmission device and the capacitive plates when using air as the dielectric medium. To simplify the simulation and reduce computational load, the outermost plates, which are farther from the resonant coils, have been optimized. Table 3 presents the mutual inductance of the radiofrequency energy transmission module and the capacitance values of the planar plates for different spacing selections, where C A B, C A C, C A D, C B C, C B D, C C D represent the equivalent capacitance values for the mutual coupling between the selected capacitor plates.

Magnetic field distribution simulations showing coupling interference between the non-contact WPT coils and capacitive plates at various spacings.
Figure 8. Magnetic field distribution simulations showing coupling interference between the non-contact WPT coils and capacitive plates at various spacings.

TABLE 3. Effect of different pole-plate spacings on coupling tolerance value.

Parameter3 mm5 mm
Mutual inductance, M(μ H)8.2566.617
Lower inner-outer plates, C *AB* (pF)2.96123.7596
Upper inner-outer plates, C *CD* (pF)2.95283.2656
Upper-lower plates (inner), C *BD* (pF)7.43514.4773
Upper-lower plates (outer), C *AC* (pF)10.72446.9972
Diagonal top-bottom plates, C *AD* (fF)544.9548597.6447
Diagonal top-bottom plates C *BC* (fF)634.4726698.6884

The analysis of the simulation results in Figure 8, in conjunction with the data from Table 3, reveals several key observations. When the distance between the capacitive radiofrequency plates is smaller than the spacing of the radiofrequency energy transmission coils, as shown in Figure 8c the data transmission plates experience the greatest interference from the coupling magnetic field. In the scenario where the capacitive radiofrequency plates and the resonant coils are coplanar, as depicted in Figure 8b, the magnetic field coupling effect on the plates at the power transmission end reaches its maximum. Conversely, when the spacing between the capacitive radiofrequency plates is greater than the magnetic coupling coils, as illustrated in Figure 8a, it is evident that the coupling magnetic field between the resonant coils is predominantly concentrated between the coils, leading to a decreased coupling effect on the surrounding capacitive radiofrequency plates. Furthermore, reducing the spacing between the transmission plates can significantly enhance the coupling between the facing plates and minimize the electric field leakage between closely spaced conductors in the same plane.

In conclusion, for short-distance non-contact power transmission applications, appropriately increasing the spacing between the wireless power transfer coils and the radio-frequency capacitors under the condition of applied magnetic shielding, as indicated by finite element simulation of magnetic field distribution, can significantly enhance the reliability of simultaneous power and data transmission in short-range non-contact systems.

3.3 Effect of applied shielding on non-contact RF plates

In the experimental setup, a shielding material is applied around the inner coil to reduce the interference caused by the resonance working magnetic field leakage of the internal radiofrequency energy transmission coil on the isolated capacitive radiofrequency data transmission module. Considering the mechanical impacts such as friction on the installation mechanism within the drilling vessel due to the magnetic barrier material, a small gap is retained on the side of the power receiving coil to minimize the additional friction loss caused by the magnetic barrier material, as illustrated in Figure 9. A finite element simulation is conducted on the simulated environment, with the relevant simulation parameters referring to the model shown in Table 1.

Influence of magnetic insulating materials on coupling between magnetic field escape and transmissible plates.
Figure 9. Influence of magnetic insulating materials on coupling between magnetic field escape and transmissible plates.

Figure 9a illustrates the severe field leakage of the unshielded coil coupling onto the plates, whereas Figure 9b depicts the constrained magnetic field distribution after introducing the nanocrystalline magnetic shielding. The 0.1 mm nanocrystalline shield μ r ≈ 10 4 forms a low-reluctance magnetic shunt path, effectively redirecting the 600 kHz fringe magnetic flux away from the capacitive RF plates. Quantitatively, this shield reduces the cross-coupling magnetic field strength by approximately 40%, significantly suppressing resonant radiation and common-mode interference voltages.

Furthermore, downhole drilling fluids can potentially introduce conductive metallic debris into the plate air gap. Electrostatic simulation indicates that non-magnetic metallic debris (volume fill ratio <5%) induces a slight coupling capacitance shift of Δ C ≤ ± 0.35 pF. Due to the differential receiving configuration and the pre-stage high-pass filter (f c = 1.2 MHZ), the DC level shift and low-frequency dielectric perturbations are effectively rejected, ensuring robust data transmission without link collapse.

4 System design

4.1 Overall design of data and power simultaneous transmission system while drilling

Based on the wireless power transmission topology depicted in Figure 2 above, the comprehensive principle of the system is presented in Figure 10. US represents the external DC power supply, and Q1, Q2, Q3, and Q4 constitute the full-bridge inverters that furnish high-frequency power to the primary resonant coil. Generally, due to the resonance deviation caused by production and installation process inaccuracies, the resonance compensation capacitances C s and C r on both sides of the mutual inductance coil should be adjusted respectively in accordance with the actual measurement of the inductance values on both sides of the mutual inductance coil.

Non-contact magnetic coupling and capacitive RF data transmission system structure.
Figure 10. Non-contact magnetic coupling and capacitive RF data transmission system structure.

In light of the results of the previous simulation analysis, to address the problem of small signal amplitude and relatively large environmental noise in data transmission via non-contact capacitive isolation, a differential amplification circuit with gain is introduced under the precondition of utilizing non-metallic - based nanocrystalline magnetic separator materials.

The overall architecture is divided into two parts: the drilling sensor side and the data receiving end. On the drilling side, the main control board acquires power from the control end and, after secondary conditioning and filtering, supplies power to the sensor transmission and passive reception conditioning circuits. It amplifies and filters the weak electrical signals obtained by the passive sensors, which are then passed through an anti-aliasing filter and into an integrated ADC for front-end data decoding. This data is then sent back to the embedded drilling control end for transmission. According to the theoretical analysis presented earlier, for the acquisition of non-contact capacitive radiofrequency signals, a differential input is adopted to amplify the signal, which includes environmental interference noise, with a lower amplification factor. An active band-pass filter circuit is then used to effectively attenuate the resonant interference noise introduced by the radiofrequency energy transmission module (600 kHz at -32 dB). An independent reference source at the receiving end provides the level for the high-speed comparator and constructs a Schmitt trigger. By setting the delay levels appropriately, the error rate of the received signal can be further reduced.

To achieve dynamic adjustment of the resonant frequency, it is necessary to implement real-time acquisition and effective value calculation of the primary voltage and current, and to estimate the frequency variation of the voltage or current within the selected acquisition time period. Therefore, a zero-crossing detection circuit is designed, which, when combined with the input capture function of an embedded chip, can rapidly compensate for the detuning caused by load changes. Figure 11 illustrates the control flow of a non-contact radiofrequency energy transmission module with resonant frequency tracking and compensation.

Resonant frequency tracking control logic.
Figure 11. Resonant frequency tracking control logic.

To ensure the reproducibility of the proposed system and to clarify the specific hardware configurations used in the experimental setup, the detailed parameters of the coils, shielding materials, semiconductor devices, and control algorithms are summarized in Table 4.

TABLE 4. Experimental setup and hardware parameters.

CategoryParameterValue/Specific model
WPT coilsWire typeSolid copper enameled wire
Wire diameter0.5 mm
Number of turns (N 1 , N 2)15/15
Resonant frequency600 kHz
Nanocrystalline shieldingThickness: 0.1 mm, μ r ≈ 10 4
Capacitive RF platesMaterialCopper
DimensionsDiameter: 30 mm, thickness: 1 mm
Circuit componentsHigh-frequency inverter (Q1-Q4)FGH40N60SMD (high-speed field stop IGBT)
Rectifier bridge (D1-D4)SS310 (Schottky diode, 3A/100 V)
Low-bias-current op-ampOPA365 (CMOS, 50 MHz bandwidth)
High-speed comparatorTLV3501 (4.5 ns high-speed comparator)
Primary/Secondary matching caps5.9 nF (C0G/NP0 ceramic capacitors)
Output filter capacitor6.8 μF
Control systemMCU model & clock speedGD32F4, 120 MHz
PI compensator coefficientsK *p* = 0.5, K *i* = 0.012
Tracking response time<2 m

4.2 Construction of test platform

Referring to the previous Figure 11, which illustrates the operational control logic of the non-contact power transmission module, and considering the summary of the design configuration in the fourth section, a verification platform for the development of electrical power and data isolation transmission in drill collars was created. The prototype consists of four printed circuit boards (PCBs) located on either side of the simulated drill collar partition. The external right side features a capacitive isolation data receiving end, which is connected to the subsequent stage of analog signal conditioning circuitry, while the left side is equipped with a data transmitting plate connected to the data transmission port of the embedded system in the drilling process. The screws at the four corners are for display purposes only and do not have electrical connections.

During normal operation, the subsequent stage of the drill pipe side’s analog signal conditioning loop is connected to an external power source, enabling non-contact power transmission to the load end; simultaneously, the embedded loop on the drill collar side pushes and pulls the upstream data for output. The proposed coaxial structure is designed to allow continuous data transmission from the drilling edge sensors during drill collar rotation; in this study, baseline functionality is validated under stationary bench conditions.

Figure 12 illustrates the design of a non-contact power and data co-transmission verification model based on the framework established in the fourth section. To validate the implementation of non-contact wireless power and data synchronous transmission, a GD32F4 MCU (operating at a 120 MHz clock speed) with an integrated DAC combined with an output amplification buffer circuit is utilized as the analog data modulation interface for upstream data. The frequency tracking control loop employs a Proportional-Integral (PI) compensator to ensure a fast response time of under 2 ms. For the receiver conditioning circuitry, low-bias-current operational amplifiers and high-speed comparators were selected to minimize signal distortion. Concurrently, an oscilloscope is used to observe the waveform of the modulated signal received by the conditioning circuitry on the simulated drill pipe side, which serves as real-time data demodulation and reception verification.

Principle test bed for non-contact electrical energy and data co-transmission system.
Figure 12. Principle test bed for non-contact electrical energy and data co-transmission system.

Figure 13a illustrates the voltage waveforms on either side of the matching capacitor at the transmitter end of the wireless power transfer module under rated load conditions. Figure 13b depicts the voltage waveforms across the resonant matching capacitor when the designed wireless power transfer module is connected to the control loop on the drilling side. Comparative analysis of simulation and actual observed waveforms reveals a positive correlation between the severity of output voltage distortion and the magnitude of current required to flow through the load. Interval recordings of the output voltage variations under the condition of connected downstream embedded load indicate that the voltage amplitude meets the requirements of the simulated drilling embedded end, as shown in Figures 13b,c.

Experimental voltage waveforms of the wireless power transfer module under different operating conditions: **(a)** transmitter capacitor voltage waveform **(b)** receiver voltage waveform with control loop load **(c)** receiver voltage waveform without load.
Figure 13. Experimental voltage waveforms of the wireless power transfer module under different operating conditions: (a) transmitter capacitor voltage waveform (b) receiver voltage waveform with control loop load (c) receiver voltage waveform without load.

4.3 Non-contact capacitive RF functional verification

The functionality of the non-contact capacitive radio-frequency data transmission link is tested, while also comparing the impact of transistor-transistor logic (TTL) level amplification buffering and 2ASK modulation transmission on the difficulty of downstream data demodulation. Figure 14a demonstrates the use of a DAC to package upstream data using the 2ASK method, and the waveform of the data at the capacitive radio-frequency receiving end before conditioning is shown.

Received signal waveforms of 2ASK modulated upstream data before conditioning: **(a)** data 0X55 **(b)** data 0XF6.
Figure 14. Received signal waveforms of 2ASK modulated upstream data before conditioning: (a) data 0X55 (b) data 0XF6.

To facilitate observation, a lower frequency division ratio is used to implement 2ASK modulation transmission of upstream data via the MCU’s integrated DAC, achieving a transmission rate of up to 66.67 kbit/s. This also verifies that when using 2ASK modulation for data packaging and uploading, the data transmission rate is positively correlated with the main frequency of the embedded device and the DAC logic conversion rate. Figure 15 shows the waveform at the output of the peak-hold circuit at the capacitive radio-frequency receiving end, after the MCU peripheral controls the output of TTL level through a buffered amplification circuit. By comparing the waveforms in Figures 14b, 15, it is evident that the level buffer amplification and peak-holding scheme significantly reduces hardware complexity while boosting throughput. In this channel, the raw differential signal amplitude is V s i g = 3.3 V peak-to-peak. The 600 kHz WPT resonant magnetic field induces an open-circuit common-mode noise of approximately V n o i s e . c m ≈ 180 mV on the plates. Combining the nanocrystalline shielding (−15 dB magnetic attenuation) and the active analog high-pass filter (−32 dB rejection at 600 kHz), the residual in-band interference is suppressed below 12 mV, achieving an operating SNR of ≈ 15 dB.

Demodulated waveforms of data 0X1A using level buffer amplification and peak-hold circuit: **(a)** Demodulated output waveform; **(b)** Peak-hold trigger waveform for FPGA comparison.
Figure 15. Demodulated waveforms of data 0X1A using level buffer amplification and peak-hold circuit: (a) Demodulated output waveform; (b) Peak-hold trigger waveform for FPGA comparison.

By optimizing the RC parameters of the peak-hold circuit and employing FPGA trigger tracking, a stable net effective payload rate of 14.4 kB/s is achieved (corresponding to an uncoded raw bit rate of 115.2 kbps, maintaining a measured BER <10−6). In terms of hardware resources, the proposed level amplification-peak holding scheme requires only an op-amp buffer and a discrete comparator, eliminating the need for high-speed DAC modulators and multi-tap digital matched filters required by 2ASK demodulation, thereby greatly simplifying downhole circuitry.

Furthermore, by using insulated conductors such as metal rods to simulate debris between capacitive radio-frequency plates, it is observed that there is no significant change in the amplitude and phase of the transmission signal, which further proves that the capacitive radio-frequency transmission system maintains robustness even in the presence of interfering media.

Table 5 presents a quantitative performance comparison between the proposed system and recent simultaneous wireless power and data transfer (SWPDT) technologies. The proposed capacitive-magnetic hybrid scheme demonstrates competitive data rates and efficiencies over a compact gap suitable for downhole applications.

TABLE 5. Comparison of simultaneous power and data transfer systems.

ReferencePower/Data carrierPower (W)Efficiency (%)Data rateGap (mm)Rotation capable
Fan et al. (2024)Inductive/Inductive5.079.019.2 kbps2.0Yes
Ye and Parspour (2024)Inductive/NFC3.380.510.0 kbps5.0No
ProposedMagnetic resonant/Capacitive RF3.081.214.4 kB/s5.0Topologically supported (static bench verified only)

4.4 Discussion on practical deployment

While Table 5 demonstrates the competitive advantages of the proposed system under static conditions, practical downhole deployment introduces dynamic environmental challenges. To evaluate rotational and axial misalignments encountered in actual drilling, electrostatic simulations were conducted for radial offsets ranging from 0 to 5 mm. The results indicate that an angular misalignment of ± 10° or an axial radial drift of 2 mm causes a coupling capacitance degradation of approximately 8.2%, which translates to a predicted SNR drop of 1.8 dB. However, since the baseline SNR is 15 dB and the Schmitt trigger provides hysteresis protection, the BER remains well within the target threshold (<10−5). Furthermore, when transitioning this prototype into field drilling environments, potential failure modes must be addressed: (1) Thermal Drifts: High downhole temperatures (up to 125 °C) can cause resonant capacitor detuning, requiring the dynamic PI zero-crossing tracking loop described in Section 3; (2) Severe Mechanical Shock: High triaxial vibrations can lead to micro-fractures in the brittle nanocrystalline ribbon, necessitating silicone elastomer potting; and (3) Conductive Mud Invasion: Ingress of high-salinity drilling mud could create low-impedance leakage paths between the capacitive rings, requiring hermetic sealing with high-temperature PEEK insulators.

5 Conclusion

This paper presents a through-wall transmission system based on non-contact capacitive radiofrequency high-rate data and magnetically coupled resonant power synchronous transmission, which can be used for power and data isolation of embedded drilling sensors, as well as for non-contact, high-efficiency transmission. The research conclusions are as follows.

  • This paper proposes a non-contact wireless power transmission and isolated data transmission structure based on capacitive radio-frequency technology. The experimental results establish the baseline feasibility of simultaneous power and data transfer in a stationary environment, laying a technical foundation towards potentially replacing traditional slip ring brushes and high-temperature batteries in future downhole applications.
  • The research presented in this paper provides a highly promising alternative to address the issues of sparking and intermittent disconnection in traditional slip ring brushes that occur due to prolonged friction during downhole drilling operations. It also involves finite element simulations to analyze the coupling between resonant magnetic fields and metal plates at various positions. By incorporating magnetic shielding materials into the system, the study demonstrates that the arrangement of wireless power transmission coils and capacitive radio-frequency plates in different planes can significantly reduce the interference from resonant magnetic fields on capacitive radio-frequency data transmission, thereby enhancing the reliability of data transmission.
  • An experimental platform was established to test the performance of parallel transmission of power and data. The results of the experiments indicate that, compared to the 2ASK modulation method, the level buffer amplification-peak holding scheme offers significant advantages in terms of hardware cost and the difficulty and reliability of data demodulation. This scheme achieved stable demodulation of data from simulated downhole embedded devices at a communication rate of 14.4 kB/s. While current experimental validations were conducted under stationary bench test conditions, demonstrating robustness against foreign objects, future work will focus on extensive dynamic testing under varying rotational speeds, vibration, and high-temperature conditions to fully validate the system’s long-term reliability as a replacement for traditional slip rings.

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Footnotes

Edited by: Yangbin Zeng, South China University of Technology, China

Reviewed by: Xinwei Song, Beijing University of Civil Engineering and Architecture, China

Kai Yan, Shaanxi University of Technology, China

Statements

Conflict of interest

The author(s) declared that this work was conducted in the absence of any commercial or financial relationships that could be construed as a potential conflict of interest.

Funding

The author(s) declared that financial support was received for this work and/or its publication. This work was financially supported by the Doctoral Research Fund of North China Institute of Aerospace Engineering (grant number BKY-2022-05), the Central Funds Guiding the Local Science and Technology Development (grant number 246Z4418G), and the Central Funds Guiding the Local Science and Technology Development entitled Research on Condition Assessment and Life Prediction of Wind Turbine Towers Based on Vibration Strain (grant number 264Z1710G).

Data availability

The original contributions presented in the study are included in the article/supplementary material, further inquiries can be directed to the corresponding author.

Author contributions

ST: Funding acquisition, Project administration, Writing – original draft, Formal Analysis, Investigation, Conceptualization, Writing – review and editing, Software, Methodology. CZ: Conceptualization, Validation, Methodology, Investigation, Software, Writing – original draft, Formal Analysis. XW: Project administration, Methodology, Investigation, Writing – original draft, Formal Analysis. PY: Methodology, Project administration, Investigation, Data curation, Validation, Supervision, Writing – review and editing. YX: Formal Analysis, Software, Methodology, Conceptualization, Project administration, Validation, Writing – review and editing, Investigation. HZ: Methodology, Conceptualization, Validation, Software, Project administration, Formal Analysis, Writing – review and editing, Investigation, Data curation.

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Citation

Tian, S., et al. (2026). Research on capacitive radio frequency non-contact communication devices under magnetic resonant power transfer. Frontiers in Electronics, Volume 7. https://doi.org/
Received
Accepted
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Volume
7 - 2026

Copyright

Copyright © 2026 Tian, Zhang, Wu, Yao, Xiang and Zhao.

License: https://creativecommons.org/licenses/by/4.0/

*Correspondence: Hanzhi Zhao

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